This invention is in the field of digital communications, and is more specifically directed to noise cancellation techniques in received digital communications.
Digital Subscriber Line (DSL) technology has become a primary technology providing high-speed Internet access, and now video and telephone communications, in the United States and around the world. As is well known in the art, DSL communications are carried out over existing telephone “wire” facilities, between individual subscribers and a central office (CO) location, operated by a telephone company or an Internet service provider. Typically, some if not all of the length of the loop between the CO and the customer premises equipment (CPE) consists of conventional twisted-pair copper telephone wire. Remarkably, modern DSL technology is able to carry out extremely high data rate communications, even over reasonably long lengths (e.g., on the order of 15,000 feet) of twisted-pair wire, and without interfering with conventional voiceband telephone communications, carried out over the twisted-pair wire simultaneously with the DSL data communications.
Modern DSL communications achieve these high data rates through the use of multicarrier modulation (MCM) techniques, more specifically discrete multitone modulation (DMT), by way of which the data signals are modulated onto orthogonal tones, or subcarriers, within a relatively wide frequency band (on the order of 1.1 MHz for conventional ADSL, and on the order of 2.2 MHz for ADSL2+), residing above the telephone voice band. The data symbols modulated onto each subchannel are encoded as points in a complex plane, according to a quadrature amplitude modulation (QAM) constellation. The number of bits of data that are carried over each subchannel (i.e., the “bit loading”), and thus the number of points in the QAM constellation for that subchannel, depend on the signal-to-noise ratio (SNR) at the subchannel frequency, which in turn depends on the noise and signal attenuation present at that frequency. For example, relatively noise-free and low attenuation subchannels may communicate data in ten-bit to fifteen-bit symbols, represented by a relatively dense QAM constellation with short distances between constellation points. On the other hand, noisy channels may be limited to only two or three bits per symbol, requiring a greater distance between adjacent points in the QAM constellation to resolve the transmitted symbol. The sum of the bit loadings over all of the subchannels in the transmission band for a DSL link of course amounts to the number of transmitted bits per DSL symbol for that link. And the data rate for DSL communications corresponds to the product of the symbol rate with the number of bits per DSL symbol.
FIG. 1 illustrates the data flow in conventional DSL communications, in a single direction (e.g., downstream, from a central office “CO” to customer premises equipment “CPE”). Typically, each DSL modem (i.e., both at the CO and also in the CPE) includes a transceiver (i.e., both a transmitter function and a receiver function), so that data is also communicated in the opposite direction over transmission channel LP according to a similar DMT process. As shown in FIG. 1, the input bitstream that is to be transmitted, typically a serial stream of binary digits in the format as produced by the data source, is applied to constellation encoder 11 in a transmitting modem 10. Constellation encoder 11 fundamentally groups the bits in the input bitstream into multiple-bit symbols that are used to modulate the DMT subchannels, with the number of bits in each symbol determined according to the bit loading assigned to its corresponding subchannel, based on the characteristics of the transmission channel as mentioned above. Encoder 11 may also include other encoding functions, such as Reed-Solomon or other forward error correction coding, trellis coding, turbo or LDPC coding, and the like. The symbols generated by constellation encoder 11 correspond to points in the appropriate modulation constellation (e.g., QAM), with each symbol associated with one of the DMT subchannels. Following constellation encoder 11, shaping function 12 derives a clip prevention signal included in the encoded signals to be modulated, to reduce the peak-to-average ratio (PAR) as transmitted as described in commonly assigned U.S. Pat. No. 6,954,505, issued Oct. 11, 2005, and incorporated herein by this reference.
These encoded symbols are applied to inverse Discrete Fourier Transform (IDFT) function 13, which associates each symbol with one subchannel in the transmission frequency band, and generates a corresponding number of time domain symbol samples according to the Fourier transform. As known in the art, cyclic insertion function 14 appends a cyclic prefix or suffix, or both, to the modulated time-domain samples from IDFT function 13, and presents the extended block of serial samples to parallel-to-serial converter 15. Cyclic insertion function 14 may follow rather than precede parallel-to-serial converter 15 in the transmission sequence, in some implementations. In either case, the time-domain serial sequence, as may be upsampled (not shown) as appropriate, is applied to digital filter function 16, which processes the datastream in the conventional manner to remove image components and voice band or ISDN interference. The filtered digital datastream signal is converted into the analog domain by digital-to-analog converter 17. After conventional analog filtering and amplification (not shown), the resulting DMT signal is transmitted over a channel LP, over some length of conventional twisted-pair wires, to a receiving DSL modem 20, which, in general, reverses the processes performed by the transmitting modem to recover the input bitstream as the transmitted communication.
At receiving DSL modem 20, analog-to-digital conversion 22 converts the filtered analog signal into the digital domain, following which conventional digital filtering function 23 is applied to augment the function of pre-conversion receiver analog filters (not shown). A time domain equalizer (TEQ) (not shown) may apply a finite impulse response (FIR) digital filter to effectively shorten the length of the impulse response of the transmission channel LP. Frame alignment function 24 receives the sequence of filtered digital samples, and arranges these samples into frames, by removing the cyclic extension from each block of samples, and by performing serial-to-parallel conversion to apply a block of samples (2N) to Discrete Fourier Transform (DFT) function 27. DFT function 27 recovers the modulating symbols at each of the subchannel frequencies, by reversing the IDFT performed by function 12 in transmission. The output of DFT function 27 is a frequency domain representation of the transmitted symbols multiplied by the frequency-domain response of the effective transmission channel. Frequency-domain equalization (FEQ) function 28 divides out the frequency-domain response of the effective channel, recovering the modulating symbols, each representable as a point in a QAM constellation. Constellation decoder function 29 then resequences the symbols into a serial bitstream, decoding any encoding that was applied in the transmission of the signal and producing an output bitstream that corresponds to the input bitstream upon which the transmission was based. This output bitstream is then forwarded to the client workstation, or to the central office network, as appropriate for the location.
As evident from the foregoing description, the data rate attained in such DSL communications is limited by the noise present on the various subchannels, because the bit loading of each subchannel depends on the signal-to-noise ratio at that subchannel frequency. As is well known in the art, the effects of “crosstalk” dominate the noise in modern DSL links. Crosstalk is, of course, noise present on one communications link that results from the coupling, to that link, of signals and noise from other communications links in the physical vicinity. So-called “near-end” crosstalk (NEXT) is crosstalk noise on received signals that originates from the same end of the cable at which the receiver is located (i.e., noise from communications in the opposite direction from the received signals), either on a neighboring cable or facility or resulting from transmissions by the same modem receiving the noise-corrupted signal. “Far-end” crosstalk (FEXT), on the other hand, is crosstalk noise coupling onto one link from other links in the physical vicinity, carrying communications in the same direction (i.e., upstream or downstream) as the received signal of concern. Indeed, crosstalk and other interference will often dominate the true signal being carried over the DSL link, at a ratio of tens of dB.
As known in the art, most telephone systems bundle multiple twisted pairs into a single “binder”, with a given binder and the wire pairs it contains disposed between the same two points. For example, a binder may carry multiple twisted pairs deployed between a central office (CO) and a neighborhood distribution frame. Of course, the close proximity of twisted pairs within a common binder is a typical source of crosstalk among the signals carried by those twisted pairs.
FIG. 2 illustrates the architecture of a conventional DSL communications system. At central office CO, DSL modem 30 communicates on one side with routers connected to Internet service providers or other servers, in the conventional manner. CO modem 30, for example if based on the AC7 ADSL Infrastructure Chipset available from Texas Instruments Incorporated, can support up to sixteen separate DSL links, four of which are shown in FIG. 2 in this example as supported by a corresponding one of DSL modem ports 350 through 353. Each DSL modem port 350 through 353 communicates with a respective customer premises equipment (CPE) modem 400 through 403, respectively, over a corresponding twisted-pair wire facility TWP0 through TWP3. In this conventional example, twisted-pair wire facilities TWP0 through TWP3 are “bonded” into a single physical binder B, with twisted-pair wire is used for the entire loop length from the CO to the CPE.
Of course, other twisted-pair wire facilities besides those supported by DSL modem 30 at the CO may also be included within the same binder B; indeed, binder B may carrier twisted pairs from other central offices. Furthermore, as well known in the art, the CO will typically also combine other signals onto the same twisted pair facilities TWP. Commonly, voice telephone signals (“plain old telephone service”, or “POTS”) are also carried by twisted-pair wire facilities TWP, at frequencies below those of the downstream (CO-to-CPE) and upstream (CPE-to-CO) DSL data communications.
By way of further background, as known in the art, the data signals communicated over twisted-pair wire facilities TWP according to conventional DSL communications are differential mode signals, in that the information is conveyed by the voltage difference between the two wires in the twisted-pair. In contrast, the common mode voltage of the wires in the twisted-pair (i.e., the absolute voltages of the wires relative to a reference voltage, or ground) does not carry information. Conventional “analog front-end” or “AFE” circuits at the interface between CO modem 30 and CPE modem 40 and the twisted-pair TWP are designed to reject common-mode “signals”, which in this application are purely noise (no information being by the common-mode voltages). Ideally, this common-mode rejection is perfect, such that the differential mode signal can be recovered regardless of the common mode voltage on the twisted-pair, and regardless of variations of that common mode over time. However, there is generally some amount of coupling of common-mode noise onto the differential mode signal in modern DSL installations. This coupling, along with the non-ideal common-mode rejection of modern AFE circuits, results in common-mode noise appearing in the differential-mode signal. In typical DSL installations, the ratio of common-mode to differential-mode in received signals is on the order of 60 dB.
It has been observed, in connection with this invention, that crosstalk interference (of both the FEXT and NEXT type) appears both as noise in the differential-mode, and also as common-mode noise. In addition, it has been observed that other interferers in DSL communications, such as narrow-band radio-frequency (RF) interferers, also appear as both common-mode and differential-mode noise.